Mobile wireless phone and data communication have become increasingly popular. These applications, however, pose two special problems. First, the RF carrier modulation by which information is transmitted must demand the smallest bandwidth possible due to the general shortage of available spectrum. As a result, both the amplitude and the phase (i.e., frequency) of the carrier must be modulated. Amplifying the modulated carrier without excessive distortion in the transmitter output stage imposes significant linearity constraints on the output stage amplifier.
Second, the power efficiency of the mobile transmitter is very important because the mobile end of wireless communication link is battery powered. Typically, the transmitter output stage is the largest power consumer; hence, improvement in this stage are the most important. One of the most efficient power amplifiers are the class C RF amplifiers in which the output transistor conducts current only at the time when the collector-emitter voltage is at its lowest value. Unfortunately, these amplifiers are very nonlinear and introduce substantial amplitude distortion. Because of this distortion, class C amplifiers are used mainly in FM transmitters in which the amplitude or "envelope" of the RF carrier is constant, and hence, such distortion has no effect.
One method for avoiding this distortion with class C amplifiers that still allows linear amplitude modulation is to generate two signals with constant amplitude using the class C amplifiers and then combine these signals. The amplitude modulation is achieved by modulating the relative phase of the two constant amplitude signals. Denote the two signals by V.sub.1 and V.sub.2, respectively. EQU V.sub.1 =V sin [.omega.t+mt(t)+a(t)] (1) EQU and EQU V.sub.2 =V sin [.omega.t+mt(t)-a(t)] (2)
These two signals are added vectorially in a "power combiner" to generate a phase and amplitude modulated carrier V.sub.out =2q(t)V sin [.omega.+m(t)], where V is the amplitude of both V.sub.1 and V.sub.2 and O&lt;q(t)&lt;1. Here m(t) and q(t) are the desired phase and amplitude modulations of the resulting carrier v, and a(t) and -a(t) are the additional phase modulations of the two constant envelope components V.sub.1 and V.sub.2. If the power combiner generates the output signal V by vectorially adding V.sub.l and V.sub.2, the resulting amplitude modulation q(t) will be q(t)=cos [a(t)]. If the power combiner vectorially subtracts V.sub.1 and V.sub.2, q(t) will be q(t)=sin [a(t)].
In all of the following discussions we will assume that the bandwidth of phase modulation m(t), amplitude modulation q(t), and phase modulation a(t), is only a small fraction of the carrier frequency.
Methods of generating the two signals V.sub.1 =V sin [.omega.t+m(t)+a(t)] and V.sub.2 =V sin [.omega.t+m(t)-a(t)] resulting in the desired signal V.sub.out =2q(t)V sin [.omega.t+m(t)] at the output of the power combiner are known to the prior art. For example, F. J. Casadevall, RF Design, February 1990, pp. 41-48 produces the components V.sub.1 and V.sub.2 starting with a low power, low frequency .omega.' version of the desired phase and amplitude modulated output signal. Casadevall generates a signal v'.sub.in =q(t)V.sub.in sin [.omega.'t+m(t)]. Here, .omega.'&lt;&lt;, the final carrier frequency. This signal is digitized and a digital signal processor is used to generate Cartesian components I.sub.1, Q.sub.1 representing V.sub.1, and I.sub.2, Q.sub.2, representing V.sub.2 but at the low frequency .omega.'. Then, V.sub.in, is upconverted with the aid of an RF oscillator, a phase shifter and mixers. This is accomplished by upconverting the two sets of I and Q components to frequency .omega.. Next, the upconvened I and Q components are summed to generate the constant envelope components V.sub.1 and V.sub.2 which are then used to drive two highly efficient power amplifiers which, in turn, drive the power combiner.
This approach has several disadvantages. First, the circuitry has a high degree of complexity. Second, any phase or amplitude errors in the mixers and in the power amplifiers appear as distortion at the power combiner output. Third, this circuit cannot simply be substituted for a conventional RF power amplifier because the input frequency must be low to match the speed of the digitizer and digital signal processor.
A second prior art solution to this problem is presented in D. C. Cox, IEEE Transactions on Communications, December 1974, pp. 1942-1945. This system generates a(t)=arc sin [q(t)] by starting with a low power, phase and amplitude modulated input signal v.sub.in =q(t)V.sub.in sin [.omega.t+m(t)] at the final frequency .omega.. This signal is processed in parallel by a limiter to generate a constant envelope, phase modulated only signal u=U sin [.omega.+m(t)], and an envelope detector to generate a baseband replica of the amplitude modulation q(t). A feedback loop which includes a phase modulator feeding a phase detector is also used. The phase modulator modulates the signal from the limiter (u=U sin [.omega.t+m(t)] by an amplified error signal causing the phase detector output to follow q(t). This implements an inverse sine function so that the signal at the input of the phase modulator, when amplified by the constant envelope amplifier, is V.sub.1 =V sin [.omega.t+m(t)+a(t)], where a(t)=arc sin [q(t)]. Signal V.sub.2 with phase angle -a(t) is derived in a similar way.
This approach has two disadvantages. First, it is not suitable for modulation schemes in which the carrier assumes temporarily zero amplitude, such as BPSK and QPSK. The limiter supplying u=U sin [.omega.t+m(t)] loses its output signal during the zero carrier amplitude period. Second, any amplitude or phase errors in the power amplifiers appear as distortion at the power combiner output.
A third prior art solution to this problem is presented in D. C. Cox and R. P. Leek, "Component Signal Separation and Recombination for Linear Amplification with Nonlinear Components", IEEE Transactions on Communications, November 1975, pp. 1281-1287. This system uses essentially the same method as the above-described Cox publication, except frequency multiplication is used. The frequency multiplication decreases the operating range, and thus, improves the linearity of the phase modulator. The phase modulator is driven by a frequency equal to one third of the operating frequency (.omega./3), its output frequency is then multiplied by 3 to .omega. in a frequency multiplier. Since frequency multiplication multiplies phase changes, the phase modulator operates over a third of the phase angle range required by a(t). Unfortunately this approach suffers from the same problems as that described with reference to Cox's original proposal.
A fourth prior art attempt to solve this problem is described in Leon Couch "A VHF Linc Amplifier", Conference Proceedings IEEE Southeastcon '82, Apr. 4-7, 1982. In this system, analog signal processing is used to develop the I and Q components of both V.sub.1 and V.sub.2 in a manner analogous to that described in the Casaderail reference. The circuits first separate the input signal into a limited phase modulated signal and a baseband envelope signal. The envelope signal is acted on by various circuitry including squaring and square rooting circuitry. This solution suffers from the same disadvantages as the systems taught by Cox. In addition, an analog squaring and an analog square rooting circuit are required which increases the system cost and complexity.
Finally, A. Bateman, "The Combined Analogue Locked Loop Universal Modulator (CALLUM)", IEEE Vehicular Technology Conference Digest, 1992, pp. 759-763 teaches a system which generates V.sub.1 and V.sub.2 from the baseband I and Q components of the input signal. In this circuit, low power versions of V.sub.1 and V.sub.2 are generated by two voltage controlled oscillators (VCOs). The frequency control input of each VCO is driven so that the correct frequency and phase are generated for V.sub.1 and V.sub.2. This is done by generating I and Q signals from the output signal obtained by adding V.sub.I and V.sub.2. These I and Q signals are compared with the original input I and Q signals. Any error between compared signals is amplified and drives the VCO frequency control input, forcing the error to zero.
The approach also has two disadvantages. First, for full 4-quadrant operation, the CALLUM modulator must be complemented by a switching matrix not described in the paper. Further, the CALLUM modulator cannot be simply substituted for conventional power amplifiers because it requires baseband I and Q inputs.
Broadly, it is the object of the present invention to provide an improved circuit for generating phase and amplitude modulated signals with high efficiency.
It is a further object of the present invention to provide a modulation circuit in which any amplitude or phase errors in the power amplifiers do not appear as distortions at the power combiner output.
It is yet another object of the present invention to provide a modulation circuit that can be simply substituted for a conventional RF power amplifier
These and other objects of the present invention will become apparent to those skilled in the art from the following detailed description of the invention and the accompanying drawings.